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Preliminary
RT8805
Two Phase General Purpose PWM Controller
General Description
The RT8805 is the most compact dual-phase synchronous buck controller in the industry specifically designed for high power density applications. This part is capable of delivering up to 60A output current due to its embedded bootstrapped drivers that support 12V + 12V driving capability. The phase currents are sensed by innovative time sharing RDS(ON) current sensing technique for current balance and over current balance. Using one common GM amplifier to sense two phase currents eliminates offset and nonlinearity of the GM amplifier and yields good current balance. Other features include adjustable operation frequency from 50kHz to 1MHz, adjustable soft-start, PGOOD, external compensation, enable/shutdown for various application and performance consideration. The RT8805 comes to a tiny footprint package of VQFN-16L 3x3 package that is capable of dissipating up to 1.47W heat.
Features
12V Power Supply Voltage 2 Phase Power Conversion Embedded 12V Boot Strapped Driver Precise Core Voltage Regulation Low Side MOSFET RDS(ON) Current Sensing for Power Stage Current Balance External Compensation Adjustable Soft-Start Adjustable Frequency and Typical at 300kHz Per Phase Power Good Indication Adjustable Over Current Protection Small 16-Lead VQFN Package RoHS Compliant and 100% Lead (Pb)-Free
Applications
Middle-High End GPU Core Power High End Desktop PC Memory Core Power Low Output Voltage, High Power Density DC-DC Converters Voltage Regulator Modules
Ordering Information
RT8805 Package Type QV : VQFN-16L 3x3 (V-Type) Operating Temperature Range P : Pb Free with Commercial Standard
Pin Configurations
(TOP VIEW)
PHASE1 LGATE1 LGATE2
12 PHASE2 11 UGATE2 10 BOOT2 9 5 6 7 8
Note : RichTek Pb-free products are : RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020. Suitable for use in SnPb or Pb-free soldering processes. 100% matte tin (Sn) plating.
UGATE1 1 BOOT1 2 AGND 3 IMAX 4
16 15 14 13
GND
VCC
PGOOD
RT
SS
Marking Information
For marking information, contact our sales representative directly or through a RichTek distributor located in your area, otherwise visit our website for detail.
VQFN-16L 3x3
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COMP
FB
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RT8805
Typical Application Circuit
Preliminary
VIN
3.3VCC 12VCC
R8 R9
9 14 C9
RT8805 2 PGOOD BOOT1 VCC UGATE1 1 PHASE1 16 LGATE1 15 10 11 12 13 Q4 C6 Q3 Q2 C4 Q1
C5
R1 R2
5 4 3 8
RT IMAX AGND FB COMP
L1
VOUT
C8 BOOT2 UGATE2
C1 C2 R3
7
C7 L2
6 EN Q5 C3
SS GND
PHASE2 LGATE2
R4
R5
Function Block Diagram
VCC PWMCP PGOOD 0.8V FB COMP External Soft Star SS CLK1 AGND RAMP1 RAMP2 CLK2 + PWMCP PWM2 Logic + EA VCC + PWM1 Logic VCC BOOT1 UGATE1 PHASE1 LGATE1 BOOT2 UGATE2 VCC PHASE2 LGATE2
Current Balance S/H GM + OCP Reg VDD CLK1 CLK2 OC
MUX
PHASE2 PHASE1
IMAX VCC
To PWM Logic Central Logic OC
PGOOD RT Clock
GND
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Preliminary Functional Pin Description
UGATE1 (Pin 1), UGATE2 (Pin 11) Upper Gate Drive. These pins drive the gates of the highside MOSFETs. BOOT1 (Pin 2), BOOT2 (Pin 10) Bootstrap Power Pin. These pins power the high-side MOSFET drivers. Connect These pins to the junctions of the bootstrap capacitors. AGND (Pin 3) Chip Analog Ground. IMAX (Pin 4) Maximum Current Setting. This pin sets the current limiting level. Connect this pin with resistor to ground to set the current limit. RT (Pin 5) Timing Resistor. Connect a resistor from RT to AGND to set the clock frequency. SS (Pin 6) Soft-Start Pin. This pin provides soft-start function for controller. The COMP voltage of the converter follows the ramping voltage on the SS pin. COMP (Pin 7) Compensation Pin. This pin is output node of the error amplifier. FB (Pin 8) Feedback Pin. This pin is negative input pin of the error amplifier. PGOOD (Pin 9) Power Good. PGOOD is an open drain output used to indicate the status of the voltages on SS pin and FB pin. PGOOD will go high impedance when SS > 3.7V and FB > 0.6V. LGATE1 (Pin 15), LGATE2 (Pin 13) Lower Gate Drive. These pins drive the gate of the lowside MOSFETs. PHASE1 (Pin 16), PHASE2 (Pin 12)
RT8805
These pins are return nodes of the high-side driver. Connect These pins to high-side MOSFET sources together with the low-side MOSFET drains and the inductors. VCC (Pin 14) The VCC pin is the external 12V power. Internal 5V power (VDD) is regulated from this pin. This pin also powers the low side MOSFETS drivers. GND (Exposed Pad) Exposed pad should be soldered to PCB board and connected to GND.
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RT8805
Absolute Maximum Ratings
Preliminary
(Note 1)
Supply Voltage, VCC -------------------------------------------------------------------------------------------------- -0.3V to 16V PHASE to GND DC ------------------------------------------------------------------------------------------------------------------------- -5V to 15V < 200ns ------------------------------------------------------------------------------------------------------------------ -10V to 30V BOOT to PHASE ------------------------------------------------------------------------------------------------------ 15V BOOT to GND DC ------------------------------------------------------------------------------------------------------------------------- -0.3V to VCC+15V < 200ns ------------------------------------------------------------------------------------------------------------------ -0.3V to 42V Input, Output or I/O Voltage ----------------------------------------------------------------------------------------- GND-0.3V to 7V Power Dissipation, PD @ TA = 25C VQFN-16L 3x3 --------------------------------------------------------------------------------------------------------- 1.47W Package Thermal Resistance (Note 4) VQFN-16L 3x3, JA --------------------------------------------------------------------------------------------------- 68C/W Junction Temperature ------------------------------------------------------------------------------------------------- 150C Lead Temperature (Soldering, 10 sec.) --------------------------------------------------------------------------- 260C ESD Susceptibility (Note 2) HBM (Human Body Mode) ------------------------------------------------------------------------------------------ 1.5kV MM (Machine Mode) -------------------------------------------------------------------------------------------------- 200V
Recommended Operating Conditions
(Note 3)
Supply Voltage --------------------------------------------------------------------------------------------------------- 9V to 14V Junction Temperature Range ---------------------------------------------------------------------------------------- -20C to 70C
Electrical Characteristics
(VIN = 12V, TA = 25C, unless otherwise specified)
Parameter Supply Input Power Supply Voltage Power On Reset Power On Reset Hysteresis Power Supply Current Soft Start Soft Start Current Oscillator Free Running Frequency Frequency Variation Frequency Range Maximum Duty Cycle fOSC RT = 33k 255 -15 50 70 300 -300 75 345 15 1000 80 kHz % kHz % ISS 8 10 15 A IVCC VSS = 0V VCC VCC -5.4 --12 5.9 0.3 10 15 6.5 --V V V mA Symbol Test Conditions Min Typ Max Units
To be continued
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Preliminary
Parameter Reference Voltage Feedback Voltage Error Amplifier DC Gain Gain-Bandwidth Product Trans-conductance MAX Current (Source & Sink) Current Sense GM Amplifier OC Gate Driver Maximum Upper Drive Source Upper Drive Sink Maximum Lower Drive Source Lower Drive Sink Protection Under Voltage Protection Power Sequence Power Good Threshold Power Good Sink Capability (4mA) 3.4 -3.7 0.05 0.55 0.6 IUGATE(MAX) BOOT - PHASE = 12V RUGATE RLGATE VUGATE = 1V VLGATE = 1V ILGATE(MAX) PVCC = 12V 1 -1 --3.5 -2 VPHASE RIMAX = 33k --220 GBW GM ICOMP CLOAD = 5pF RLOAD = 20k VCOMP = 2.5V 60 6 600 300 70 10 660 360 VFB VFB = 0.8V 0.784 0.8 Symbol Test Conditions Min Typ
RT8805
Max 0.816 Units V
-----
dB MHz A/V A
--
mV
-7 -4
A A
0.65
V
4 0.2
V V
Note 1. Stresses listed as the above "Absolute Maximum Ratings" may cause permanent damage to the device. These are for stress ratings. Functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may remain possibility to affect device reliability. Note 2. Devices are ESD sensitive. Handling precaution recommended. Note 3. The device is not guaranteed to function outside its operating conditions. Note 4. JA is measured in the natural convection at T A = 25C on a low effective thermal conductivity test board of JEDEC 51-3 thermal measurement standard.
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RT8805
Preliminary
Typical Operating Characteristics
Phase Loading vs. Output Loading
30 25
VREF vs. Temperature
0.7945 0.794 0.7935 0.793
Low-Side : IPD06N03 High-Side : IPD09N03
Phase Loading (A)
20 15
V REF (V)
40 45 50
PHASE2
0.7925 0.792 0.7915 0.791 0.7905 0.79
PHASE1
10 5 0 5 10 15 20 25 30 35
0.7895 -40 -25 -10 5 20 35 50 65 80 95 110 125
Output Loading (A)
Temperature (C)
FOSC vs. Temperature
306 304 302 300 298 296 294 -40 -25 -10 5 20 35 50 65 80 95 110 125
Dead Time
RRT = 33k Low-Side : IPD06N03 High-Side : IPD09N03 UGATE No Load
F OSC (kHz)
PHASE LGATE UGATE-PHASE
(5V/Div)
Temperature (C)
Time (100ns/Div)
OCP
OCP
Start up then Short, CSS = 0.1F
VOUT (100mV/Div)
VOUT (1V/Div) SS (5V/Div)
IL (10A/Div) SS (2V/Div)
IL (20A/Div) UGATE (20V/Div) Time (25ms/Div)
Short then Start up, CSS = 0.1F
Time (25ms/Div)
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Preliminary
RT8805
Power On
No Load
Power Off
IOUT = 3A
VOUT (1V/Div) UGATE (20V/Div) LGATE (10V/Div) IL (5A/Div) Time (100s/Div)
VOUT (1V/Div) UGATE (20V/Div) LGATE (10V/Div) IL (5A/Div) Time (1ms/Div)
Short Pulse
Low-Side : IPD06N03 High-Side : IPD09N03 During Soft Start No Load
Shutdown by SS Pin
UGATE UGATE-PHASE PHASE LGATE
VOUT (500mV/Div) UGATE (10V/Div) LGATE (10V/Div)
(5V/Div)
Time (100ns/Div)
Time (100s/Div)
Start Up by SS Pin
No Load, CSS = 0.1F VIN = 0V, CSS = 0.1F
UVP
VOUT (1V/Div) LGATE (10V/Div) UGATE (10V/Div) SS (1V/Div) Time (5ms/Div)
VOUT (20mV/Div) SS (2V/Div) LGATE (10V/Div) UGATE (1V/Div) Time (50ms/Div)
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RT8805
Applications Information
Power On Reset
Preliminary
Frequency setting The converter switching frequency is programmed by connecting a resistor from the RT pin to GND. Figure 2 illustrates switching frequency vs. RRT.
Switching Frequency (kHz)
RT8805 operates with input voltage at VCC pin ranging from 5.9V to 15V. An internal linear regulator regulates the input voltage to 5V for internal control circuit use. The POR (power on reset) circuitry monitors the supply voltage to make sure the supply voltage is high enough for RT8805 normal work. When the regulated power exceeds 4.2V typically, the RT8805 releases the reset state and works according to the setting. Once the regulated voltage is lower than 4.0V, POR circuitry resets the chip. Hysteresis between the rising and falling thresholds assure that once enabled, the RT8805 will not inadvertently turn off unless the bias voltage drops substantially (see Electrical Specifications). Enable, Soft Start and Power Good Once POR releases, the RT8805 begins its soft start cycle as shown in Figure 1. A 10A source current charges the capacitor CSS connected to SS to control the soft start behavior of RT8805. During soft start, SS voltage increases linearly and clamps the error amplifier output. Duty cycle and output voltage increase accordingly. The soft start limits inrush current from input capacitors. The RT8805 regards SS pin voltage higher than 3.7V as the end of soft start cycle. Then RT8805 trip PGOOD to high impedance if no fault occurs indicating power good. The SS pin also act as the timer during OCP and UVP hiccup as described in the later sections.
Switching Frequency vs. RT Resistance
1200 1000 800 600 400 200 0 0 20 40 60 80 100
RT Resistance (k ) (k)
Figure 2. Switching Frequency vs. RRT. Voltage Control The voltage control loop consists of error amplifier, multiphase pulse width modulator, drivers and power components. As conventional voltage mode PWM controller, the output voltage is locked at the positive input of error amplifier and the error signal is used as the control signal of pulse width modulator. The PWM signals of different channels are generated by comparison of EA output and split-phase sawtooth wave. Power stage transforms VIN to output by PWM signal on-time ratio. Current Sensing Setting RT8805 senses the current of low side MOSFET in each synchronous rectifier when it is conducting for channel current balance and OCP detecting. The multiplexer and sensing GM amplifier converts the voltage on the sense component (can be a sense resistor or the RDS(ON) of the low side MOSFET) to current signal into internal circuit (see Figure 3).
VDD POR SS SSH FB PGOOD > 0.6V > 3.7V
Figure 1. Power Sequence
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Preliminary
Current Balance
PHASE1 PHASE2 CLK1 RAMP1 RAMP2 CLK2 MUX
RT8805
Current Balance S/H OCP OC
GM +
RT8805 senses the voltage drop of the low-side MOS and translates this to control the ramp signal. We can see that the voltage signal finally injected to channel one is proportional to (IL1 - IL2). Channel two is proportional to (IL2 - IL1). In steady state and current balance situation, there is no sensed signal injected into the ramp. If IL1 > IL2, the ramp bottom of channel 1 will be lifted up and decreased the duty of UGATE1. On the other hand, the ramp bottom of channel 2 will be pulled low to increase the duty of UGATE2. Finally, the loop will be back to the balance state through above mentioned negative feedback scheme. Figure 5 shows this scheme.
VREF RAMP2 + + COMP Logic & Driver k2 = k x RON2 -1 2 VIN2 IL2 L2 VON2 CL VIN1 + RAMP1 Logic & Driver k1 = k x RON1 IL1 L1 VON1 VOUT
IMAX
Figure 3. Current Sensing Loop The sensing circuit gets IX = IL(S/H) x RDS(ON) x GM by local feedback. IX is sampled and held just before low side MOSFET turns off (See Figure 4). Therefore, IX(S/H) = IL(S/H) x RDS(ON) x GM
IL(S/H) = IL(AVG) - VOUT x TOFF , L 2 TOFF = VIN - VOUT x 5 s, VIN
VCSO2 = k2 x IL2 = k x VON2 RL
FSW = 200kHz VOUT x VIN - VOUT x 5 s VIN I X (S/H) = IL(AVG) - 2L x RDS(ON) x GM
VCSO1 = k1 x IL1 = k x V ON1
Figure 5. Current Balance
Falling Slope = VOUT/L IL IL(AVG) Inductor Current IL(S/H)
Gate control a. Before SS signal reach the valley of the ramp voltage, UGATE and LGATE will be off. b. If SS pin is pulled down 0.4V, UGATE and LGATE will be off. c. UV protect function caused by FB < 0.6V and SS > 3.7V, and controller will trigger Always Hiccup Mode. d. When OC function occurs and SS > 3.7V, a constant current of 10A starts to discharge the capacitor connected to SS pin right away. When OC occurs, UGATE and LGATE will be off. When the voltage at the capacitor connected to SS pin pass about 0.4V, a constant current of 10A starts to charge the capacitor. The PWM signal is enable to pass to UGATE and
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High Side MOSFET Gate Signal
Low Side MOSFET Gate Signal
Figure 4. Inductor Current and Gate signals
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RT8805
Preliminary
The first step is to calculate the complex conjugate poles contributed by the LC output filter. The output LC filter introduces a double pole, 40dB/decade gain slope above its corner resonant frequency, and a total phase lag of 180 degrees. The resonant frequency of the LC filter expressed as follows:
FP(LC) = 1 2 x L OUT x COUT
LGATE. OCP function monitors both channels, either one can activate OCP. If the OC protection occurs three times, OCSD (Over Current Shut Down) will be activated and shut down the chip. e. When fault conditions occur or SS < 0.4V, the current sense function will be disabled. Power Good PGOOD goes high when soft-start voltage > 3.7V, and no fault conditions. Feedback Loop Compensation The RT8805 is a voltage mode controller ; the control loop is a single voltage feedback path including an error amplifier and PWM comparator. In order to achieve fast transient response and accurate output regulation, an adequate compensator design is necessary. The goal of the compensation network is to provide adequate phase margin (greater than 45 degrees) and the highest 0dB crossing frequency. To manipulate loop frequency response under its gain crosses over 0dB at a slope of -20dB/ decade. 1) Modulator Frequency Equations RT8805 is a voltage mode buck converter using the high gain error amplifier with transconductance (OTA, Operational Transconductance Amplifier), as Figure 6 shown. The Transconductance:
GM = IOUT VM VM = (EA+) - (EA-) ; IOUT = E/A output current.
VOUT EA+ EA+ GM ROUT
The next step of compensation design is to calculate the ESR zero. The ESR zero is contributed by the ESR associated with the output capacitance. Note that this requires that the output capacitor should have enough ESR to satisfy stability requirements. The ESR zero of the output capacitor expressed as follows: 1 FZ(ESR) = 2 x COUT x ESR 2) Compensation Frequency Equations The compensation network consists of the error amplifier and the impedance networks ZC and ZF as Figure 7 shown.
R1 VREF + GM VCOMP C2 R2 C1
VOUT FB
RF
Figure 7. Compensation Loop
FZ1 = FP1 = FP2 =
1 2 x R2 x C2 1 2 x R1x C1 1 2 x R2 x C1x C2 C1+ C2
Figure 6. OTA Topology This transfer function of OTA is dominated by a higher DC gain and the output filter (LOUT and COUT) with a double pole frequency at FLC and a zero at FESR. The DC gain of the modulator is the input voltage (VIN) divided by the peak to peak oscillator voltage VRAMP.
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Figure 8 shows the DC-DC converter's gain vs. frequency. The compensation gain uses external impedance networks ZC and ZF to provide a stable, high bandwidth loop. High crossover frequency is desirable for fast transient response, but often jeopardize the system stability. In order to cancel one of the LC filter poles, place FZ1 before the LC filter resonant frequency. In the experience, place FZ1 at 10% LC filter resonant frequency. Crossover frequency should be higher than the ESR zero but less than 1/5 of the switching frequency. The FP2 should be place at half the switching frequency.
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Preliminary
80 80 Loop Gain 60 40 40 20 Gain (dB) 0 -20
-4040 0
RT8805
Type 3 will induce three poles and two zeros. Zeros :
Compensation Gain
FZ1 = FZ2 =
1 2 x R2 x C2 1 2 x (R1+ R3) x C3
Modulator Gain
Poles :
FP1 =
10z 0H v b c m 2100l ) d(op) vb o d(
-6060 1H 0z 10db(vo) v
1k 10k Feuny rqec Frequency (Hz)
10H .Kz
1Kz 0H
100k
10H 0Kz
10H .Mz
1M
1 2 x R2 x C1x C2 C1+ C2 1 2 x R3 x C3 1 ; R1x R3 x C1 2 x R1+ R3
FP2 = FP3 =
Figure 8. Type 2 Bode Plot There is another type of compensation called Type 3 compensation that adds a pole-zero pair to the Type 2 network. It's used to compensate output capacitor whose ESR value is much lower (pure MLCC or OSCON Capacitors). As shown in Figure 9, to insert a network between VOUT and FB in the original Type 2 compensation network can result in Type 3 compensation. Figure 10 shows the difference of their AC response. Type 3 compensation has an additional pole-zero pair that causes a gain boost at the flat gain region. But the gain boosted is limited by the ratio (R1+R4)/R4; if R3 << R4.
C3 R1 R3
which is in the origin. We recommend FZ1 placed in 0.5 x FP(LC); FZ2 placed in FP(LC); FP1 placed in FESR and FP2 placed in 0.5 x FSW. Figure 11 shows Type 3 Bode Plot.
Loop Gain
60 40 20 0
Compensation Gain
dB
Gain
-20 -40
VOUT
FB
+ GM -
VCOMP C2 R2 C1
Modulator Gain
-60 -80 2 3 4 5 6 7
R4
Log Frequency
Figure 9. Additional Network of Type 3 Compensation (Add between VOUT and FB)
Figure 11. Type 3 Bode Plot
Protection
FP3 Add Type 3 compensation FP1 FZ1 FZ2 Pole FP2
OCP The RT8805 uses " Cycle by Cycle" current comparison. The over current level is set by IMAX pin. When OC function occurs and SS > 3.7V, a constant current of 10A starts to discharge the capacitor connected to SS pin right away. When OC occurs, UGATE and LGATE will be off.
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Original Type 3 compensation
Figure 10. AC Response Curves of Type 2 and 3
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RT8805
Preliminary
UVP
VIN = 0V
When the voltage at the capacitor connected to SS pin pass about 0.4V, a constant current of 10A starts to charge the capacitor. The PWM signal is enabled to pass to the UGATE and LGATE. OCP function monitors both channels, either one can activate OCP. If the OC protection occurs three times, the chip will shut down and the state will only be released by POR. RT8805 uses an external resistor R IMAX to set a programmable over current trip point. OCP comparator compares each inductor current with this reference current. RT8805 uses hiccup mode to eliminate fault detection of OCP or reduce output current when output is shorted to ground. The OCP comparator compares the difference between IX and IIMAX.
OCP Comparator IIMAX IX + -
VOUT (20mV/Div) SS (2V/Div) LGATE (10V/Div) UGATE (1V/Div) Time (50ms/Div)
Figure 12. UVP (Always Hiccup Mode) OTP Monitor the temperature near the driver part within the chip. Shutdown the chip when OTP (Typical trip point : 170C). General Design Guide This design guide is intended to provide a high-level explanation of the steps necessary to create a multi-phase power converter. It is assumed that the reader is familiar with many of the basic skills and techniques referenced below. Power Stages Designing a multi-phase converter is to determine the number of phases. This determination depends heavily on the cost analysis which in turn depends on system constraints that differ from one design to the next. Principally, the designer will be concerned with whether components can be mounted on both sides of the circuit board, whether through-hole components are permitted, the total board space available for power-supply circuitry, and the maximum amount of load current. Generally speaking, the most economical solutions are those in which each phase handles between 20 to 25 A (One Upper and one Lower MOSFET). All surface-mount designs will tend toward the lower end of this current range. If through-hole MOSFETs and inductors can be used, higher per-phase currents are possible. In cases where
DS8805-01 November 2005
For example: From Electrical Specifications : RIMAX = 33k VPHASE = -220mV Assume Low side MOSFET RDS(ON) = 3m. Get the OCP setting current is 220mV =73A per PHASE 3m (the valley of inductor's current). Change the setting current which you want from 73A per PHASE to 50A per PHASE. Following below steps: 1. Calculate phase voltage. If Low side MOSFET RDS(ON) = 3m, VPHASE_new = -150mV.
2. RIMAX_new = -220mV x 33k VPHASE_new
RIMAX_new = 48.4k
UVP By detecting voltage at FB pin when SS > 3.7V. If FB < 0.6V, the chip will trigger the always Hiccup mode and a constant current source 10A starts to charge capacitor at SS pin when SS pass 0.4V and discharge Css when SS > 3.7V. As Figure 12 shown.
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Preliminary
board space is the limiting constraint, current can be pushed as high as 40A per phase, but these designs require heat sinks and forced air to cool the MOSFETs, inductors and heat dissipating surfaces. MOSFETs The choice of MOSFETs depends on the current each MOSFET will be required to conduct, the switching frequency, the capability of the MOSFETs to dissipate heat, and the availability and nature of heat sinking and air flow. Package Power Dissipation When choosing MOSFETs it is important to consider the amount of power being dissipated in the integrated drivers located in the controller. Since there are a total of two drivers in the controller package, the total power dissipated by both drivers must be less than the maximum allowable power dissipation for the VQFN package. Calculating the power dissipation in the drivers for a desired application is critical to ensure safe operation. Exceeding the maximum allowable power dissipation level will push the IC beyond the maximum recommended operating junction temperature of 125C. The maximum allowable IC power dissipation for the 3x3 VQFN package is approximately 1.47W at room temperature. According below equations at two phases operation, it's clear to describe that the junction temperature of the chip is directly proportional to the total CISS (including CUGATE and CLGATE) of all external MOSFETs. PD = ( CUGATE x VBOOT-PHASE2 x f ) + ( CLGATE x VCC2 x f ) +
RT8805
operated under or over maximum (~125C) operation rating. Layout Considerations Layout is very important in high frequency switching converter design. If designed improperly, the PCB could radiate excessive noise and contribute to the converter instability. First, place the PWM power stage components. Mount all the power components and connections in the top layer with wide copper areas. The MOSFETs of Buck, inductor, and output capacitor should be as close to each other as possible. This can reduce the radiation of EMI due to the high frequency current loop. If the output capacitors are placed in parallel to reduce the ESR of capacitor, equal sharing ripple current should be considered. Place the input capacitor directly to the drain of high-side MOSFET. In multi-layer PCB, use one layer as power ground and have a separate control signal ground as the reference of the all signal. To avoid the signal ground is effect by noise and have best load regulation, it should be connected to the ground terminal of output. Furthermore, follows below guidelines can get better performance of IC : 1. A multi-layer printed circuit board is recommended. 2. Use a middle layer of the PC board as a ground plane and making all critical component ground connections through vias to this layer. 3. Use another solid layer as a power plane and break this plane into smaller islands of common voltage levels. 4. Keep the metal running from the PHASE terminal to the output inductor short. 5. Use copper filled polygons on the top and bottom circuit layers for the phase node. 6. The small signal wiring traces from the LGATE and UGATE pins to the MOSFET gates should be kept short and wide enough to easily handle the several Amperes of drive current. 7. The critical small signal components include any bypass capacitors, feedback components, and compensation components. Position those components close to their pins with a local GND connection, or via directly to the ground plane.
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TJ = TA + ( JA x PD ) ( is the minor factor and could be ignored) For example, according to the application we evaluated on board, the CUGATE = 1nF, CLGATE = 5nF (dual MOSFETs in parallel), VCC = 12V, VBOOT-PHASE = 12V, and operation frequency = 300kHz. PD 1nF x 122 x 300kHz + 2 x 5nF x 122 x 300kHz = 475mW / PHASE TJ = 30C+ 68C/W x 0.475W x 2 = 94.6C That means the junction temperature is most likely to be
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RT8805
Preliminary
8. RT and RIMAX resistors should be near the RT and RIMAX pin respectively, and their GND return should be short, and kept away from the noisy MOSFET GND. 9. Place the compensation components close to the FB and COMP pins. 10. The feedback resistors for both regulators should also be located as close as possible to the relevant FB pin with vias tied straight to the ground plane as required. 11. Minimize the length of the connections between the input capacitors, CIN and the power switches by placing them nearby. 12. Position both the ceramic and bulk input capacitors as close to the upper MOSFET drain as possible, and make the GND returns (From the source of lower MOSFET to VIN, CVIN, GND) short. 13. Position the output inductor and output capacitors between the upper MOSFET and lower MOSFET and the load. 14. AGND should be on the clearer plane, and kept away from the noisy MOSFET GND.
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Preliminary Outline Dimension
SEE DETAIL A L
1
RT8805
D
D2
E
E2
1
1 2
e A A1 A3
b
2
DETAIL A Pin #1 ID and Tie Bar Mark Options Note : The configuration of the Pin #1 identifier is optional, but must be located within the zone indicated.
Symbol A A1 A3 b D D2 E E2 e L
Dimensions In Millimeters Min 0.800 0.000 0.175 0.180 2.950 1.300 2.950 1.300 0.500 0.350 0.450 Max 1.000 0.050 0.250 0.300 3.050 1.750 3.050 1.750
Dimensions In Inches Min 0.031 0.000 0.007 0.007 0.116 0.051 0.116 0.051 0.020 0.014 0.018 Max 0.039 0.002 0.010 0.012 0.120 0.069 0.120 0.069
V-Type 16L QFN 3x3 Package
RICHTEK TECHNOLOGY CORP.
Headquarter 5F, No. 20, Taiyuen Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Fax: (8863)5526611
RICHTEK TECHNOLOGY CORP.
Taipei Office (Marketing) 8F-1, No. 137, Lane 235, Paochiao Road, Hsintien City Taipei County, Taiwan, R.O.C. Tel: (8862)89191466 Fax: (8862)89191465 Email: marketing@richtek.com
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